Switch Circuit and Method of Switching Radio Frequency Signals

ABSTRACT

An RF switch circuit and method for switching RF signals that may be fabricated using common integrated circuit materials such as silicon, particularly using insulating substrate technologies. The RF switch includes switching and shunting transistor groupings to alternatively couple RF input signals to a common RF node, each controlled by a switching control voltage (SW) or its inverse (SW_), which are approximately symmetrical about ground. The transistor groupings each comprise one or more insulating gate FET transistors connected together in a “stacked” series channel configuration, which increases the breakdown voltage across the series connected transistors and improves RF switch compression. A fully integrated RF switch is described including control logic and a negative voltage generator with the RF switch elements. In one embodiment, the fully integrated RF switch includes an oscillator, a charge pump, CMOS logic circuitry, level-shifting and voltage divider circuits, and an RF buffer circuit.

CROSS-REFERENCE TO RELATED APPLICATIONS Claims of Priority

This is a continuation of application Ser. No. 12/980,161, filed Dec.28, 2010, which is a continuation of application Ser. No. 12/315,395,filed Dec. 1, 2008, which issued Dec. 28, 2010 as U.S. Pat. No.7,860,499, which is a continuation of application Ser. No. 11/582,206,filed Oct. 16, 2006, now U.S. Pat. No. 7,460,852 issued Dec. 2, 2008,which is a continuation of application Ser. No. 10/922,135, filed Aug.18, 2004, now U.S. Pat. No. 7,123,898 issued Oct. 17, 2006, which is acontinuation of application Ser. No. 10/267,531, filed Oct. 8, 2002, nowU.S. Pat. No. 6,804,502 issued Oct. 12, 2004, which claims the benefitunder 35 USC 119(e) of U.S. Provisional Application No. 60/328,353,filed Oct. 10, 2001, and each of these pending applications and issuedpatents are hereby incorporated by reference herein in their entirety.

BACKGROUND

1. Field

The present disclosure relates to switches, and particularly to a switchcircuit and method of switching radio frequency (RF) signals within anintegrated circuit. In one embodiment, the switch circuit comprises CMOSdevices implemented on a silicon-on-insulator (SOI) substrate, for usein RF applications such as wireless communications, satellites, andcable television.

2. Description of Related Art

As is well known, radio frequency (RF) switches are important buildingblocks in many wireless communication systems. RF switches are found inmany different communications devices such as cellular telephones,wireless pagers, wireless infrastructure equipment, satellitecommunications equipment, and cable television equipment. As is wellknown, the performance of RF switches is controlled by three primaryoperating performance parameters: insertion loss, switch isolation, andthe “1 dB compression point.” These three performance parameters aretightly coupled, and any one parameter can be emphasized in the designof RF switch components at the expense of others. A fourth performanceparameter that is occasionally considered in the design of RF switchesis commonly referred to as the switching time or switching speed(defined as the time required to turn one side of a switch on and turnthe other side off). Other characteristics that are important in RFswitch design include case and degree (or level) of integration of theRF switch, complexity, yield, return loss and cost of manufacture.

These RF switch performance parameters can be more readily describedwith reference to a prior art RF switch design shown in the simplifiedcircuit schematics of FIGS. 1 a-1 c. FIG. 1 a shows a simplified circuitdiagram of a prior art single pole, single throw (SPST) RF switch 10.The prior art SPST switch 10 includes a switching transistor M1 5 and ashunting transistor M2 7. Referring now to FIG. 1 a, depending upon thestate of the control voltages of the two MOSFET transistors M1 5 and M27 (i.e., depending upon the DC bias applied to the gate inputs of theMOSFET switching and shunting transistors, M1 and M2, respectively), RFsignals are either routed from an RF input node 1 to an RF output node3, or shunted to ground through the shunting transistor M2 7. Actualvalues of the DC bias voltages depend upon the polarity and thresholdsof the MOSFET transistors M1 5 and M2 7. Resistor R0 9, in series withthe RF source signal, isolates the bias from the source signal and isessential for optimal switch performance. FIG. 1 b shows the “on” stateof the RF switch 10 of FIG. 1 a (i.e., FIG. 1 b shows the equivalentsmall-signal values of the transistors M1 and M2 when the RF switch 10is “on”, with switching transistor M1 5 on, and shunting transistor M2 7off). FIG. 1 c shows the “off” state of the switch 10 of FIG. 1 a (i.e.,FIG. 1 c shows the equivalent small-signal values of the transistors M1and M2 when the RF switch 10 is “off”, with switching transistor M1 5off, and shunting transistor M2 7 on).

As shown in FIG. 1 b, when the RF switch 10 is on, the switchingtransistor M1 5 is primarily resistive while the shunting transistor M27 is primarily capacitive. The “insertion loss” of the RF switch 10 isdetermined from the difference between the maximum available power atthe input node 1 and the power that is delivered to a load 11 at theoutput node 3. At low frequencies, any power lost is due to the finiteon resistance “r” 13 of the switching transistor M1 5 when the switch 10is on (see FIG. 1 b). The on resistance r 13 (FIG. 1 b) typically ismuch less than the source resistor R0 9. The insertion loss, “IL”, cantherefore be characterized in accordance with Equation 1 shown below:

IL is approximately equal to: 10r/R0 ln(10)=0.087r (in dB).  Equation 1:

Thus, at low frequencies, a 3-Ω value for r results in approximately0.25 dB insertion loss.

Because insertion loss depends greatly upon the on resistances of the RFswitch transmitters, lowering the transistor on resistances and reducingthe parasitic substrate resistances can achieve improvements ininsertion loss.

In general, the input-to-output isolation (or more simply, the switchisolation) of an RF switch is determined by measuring the amount ofpower that “bleeds” from the input port into the output port when thetransistor connecting the two ports is off. The isolation characteristicmeasures how well the RF switch turns off (i.e., how well the switchblocks the input signal from the output). More specifically, andreferring now to the “off” state of the RF switch 10 of FIG. 1 c, theswitching transistor M1 5 off state acts to block the input 1 from theoutput 3. The shunting transistor M2 7 also serves to increase theinput-to-output isolation of the switch 10.

When turned off (i.e., when the RF switch 10 and the switchingtransistor M1 5 are turned off), M1 5 is primarily capacitive with“feedthrough” (i.e., passing of the RF input signal from the input node1 to the output node 3) of the input signal determined by theseries/parallel values of the capacitors CGD off 15 (Gate-to-DrainCapacitance when the switching transistor M1 is turned off), CGS off 17(Gate-to-Source Capacitance when the switching transistor M1 is turnedoff), and CDS1 19 (Drain-to-Source capacitance when the transistor M1 isturned off). Feedthrough of the input signal is undesirable and isdirectly related to the input-to-output isolation of the RF switch 10.The shunting transistor M2 7 is used to reduce the magnitude of thefeedthrough and thereby increase the isolation characteristic of the RFswitch.

The shunting transistor M2 7 of FIG. 1 c is turned on when the switchingtransistor M1 5 is turned off. In this condition, the shuntingtransistor M2 7 acts primarily as a resistor having a value of r. Bydesign, the value of r is much less than the characteristic impedance ofthe RF source. Consequently, r greatly reduces the voltage at the inputof the switching transistor M1 5. When the value of r is much less thanthe source resistance R0 9 and the feedthrough capacitive resistance ofthe shunting transistor M2 7, isolation is easily calculated. Switchisolation for the off state of the RF switch 10 is determined as thedifference between the maximum available power at the input to the powerat the output.

In addition to RF switch insertion loss and isolation, another importantRF switch performance characteristic is the ability to handle largeinput power when the switch is turned on to ensure that insertion lossis not a function of power at a fixed frequency. Many applicationsrequire that the switch does not distort power transmitted through a“switched-on” switch. For example, if two closely spaced tones areconcurrently passed through an RF switch, nonlinearities in the switchcan produce inter-modulation (IM) and can thereby create a false tone inadjacent channels. If these adjacent channels are reserved, forinstance, for information signals, power in these false tones must bemaintained as small as possible. The switch compression, or “1 dBcompression point” (“P1 dB”), is indicative of the switch's ability tohandle power. The P1 dB is defined as the input power at which theinsertion loss has increased by 1 dB from its low-power value. Or statedin another way, the 1 dB compression point is a measure of the amount ofpower that can be input to the RF switch at the input port before theoutput power deviates from a linear relationship with the input power by1 dB.

Switch compression occurs in one of two ways. To understand how switchcompression occurs, operation of the MOSFET transistors shown in the RFswitch 10 of FIGS. 1 a-1 c are described. As is well known in thetransistor design arts, MOSFETs require a gate-to-source bias thatexceeds a threshold voltage, V_(i), to turn on. Similarly, thegate-to-source bias must be less than V_(i) for the switch to be off.V_(i) is positive for “type-N” MOSFETs and negative for “type-P”MOSFETs. Type-N MOSFETs were chosen for the RF switch 10 of FIGS. 1 a-1c. The source of a type-N MOSFET is the node with the lowest potential.

Referring again to FIG. 1 c, if a transient voltage on the shuntingtransistor M2 7 results in turning on the shunting transistor M2 7during part of an input signal cycle, input power will be routed toground and lost to the output. This loss of power increases forincreased input power (i.e., input signals of increased power), andthereby causes a first type of compression. The 1 dB compression pointin the RF switch 10 is determined by the signal swing on the input atwhich point the turned-off shunting transistor M2 7 is unable to remainoff. Eventually, a negative swing of the input falls below the potentialof the M2 gate, as well as below ground (thus becoming the source). Whenthis difference becomes equal to V_(i), the transistor M2 7 begins toturn on and compression begins. This first type of compression is causedby the phenomenon of the turning on of a normally off gate in the shuntleg of the RF switch. Once the shunting transistor M2 7 turns on, powerat the output node 3 no longer follows power at the switch input in alinear manner. A second type of RF switch compression occurs when thesource and drain of the shunting transistor M2 7 break down at excessivevoltages. For submicron silicon-on-insulator (SOI) devices, this voltagemay be approximately only +1 VDC above the supply voltage. At breakdown,the shunt device begins to heavily conduct current thereby reducing thepower available at the output.

FIG. 2 shows a simplified schematic of a prior art single pole doublethrow (SPDT) RF switch 20. As shown in FIG. 2, the prior art RF switch20 minimally includes four MOSFET transistors 23, 24, 27 and 28. Thetransistors 23 and 24 act as “pass” or “switching” transistors (similarto the switching MOSFET transistor M1 5 of FIGS. 1 a-1 c), and areconfigured to alternatively couple their associated and respective RFinput nodes to a common RF node 25. For example, when enabled (orswitched “on”), the switching transistor 23 couples a first RF inputsignal “RF₁”, input to a first RF input node 21, to the RF common node25. Similarly, when enabled, the switching transistor 24 couples asecond RF input signal “RF₂”, input to a second RF input node 22, to theRF common node 25. The shunting transistors, 27 and 28, when enabled,act to alternatively shunt their associated and respective RF inputnodes to ground when their associated RF input nodes are uncoupled fromthe RF common node 25 (i.e., when the switching transistor (23 or 24)connected to the associated input node is turned off).

As shown in FIG. 2, two control voltages are used to control theoperation of the prior art RF switch. The control voltages, labeled“SW”, and its inverse “SW_”, control the operation of the transistors23, 24, 27 and 28. The control voltages are arranged to alternativelyenable (turn on) and disable (turn off) selective transistor pairs. Forexample, as shown in FIG. 2, when SW is on (in some embodiments this isdetermined by the control voltage SW being set to a logical “high”voltage level, e.g., “+Vdd”), the switching transistor 23 is enabled,and its associated shunting transistor 28 is also enabled. However,because the inverse of SW, SW_, controls the operation of the secondswitching transistor 24, and its associated shunting transistor 27, andthe control signal SW_ is off during the time period that SW is on (insome embodiments this is determined by SW_ being set to a −Vdd value),those two transistors are disabled, or turned off, during this same timeperiod. In this state (SW “on” and SW_“off”), the RF₁ input signal iscoupled to the RF common port 25 (through the enabled switchingtransistor 23). Because the second switching transistor 24 is turnedoff, the RF₂ input signal is blocked from the RF common port 25.Moreover, the RF₂ input signal is further isolated from the RF commonport 25 because it is shunted to ground through the enabled shuntingtransistor 28. As those skilled in the transistor designs arts shalleasily recognize, the RF₂ signal is coupled to the RF common port 25(and the RF₁, signal is blocked and shunted to ground) in a similarmanner when the SW control signal is “off” (and SW_ is “on”).

With varying performance results, RF switches, such as the SPDT RFswitch 20 of FIG. 2, have heretofore been implemented in differentcomponent technologies, including bulkcomplementary-metal-oxide-semiconductor (CMOS) and gallium-arsenide(GaAs) technologies. In fact, most high performance high-frequencyswitches use GaAs technology. The prior art RF switch implementationsattempt to improve the RF switch performance characteristics describedabove, however, they do so with mixed results and with varying degreesof integrated circuit complexity and yields. For example, bulk CMOS RFswitches disadvantageously exhibit high insertion loss, low compression,and poor linearity performance characteristics. In contrast, due to thesemi-insulating nature of GaAs material, parasitic substrate resistancescan be greatly reduced thereby reducing RF switch insertion loss.Similarly, the semi-insulating GaAs substrate improves switch isolation.

Although GaAs RF switch implementations offer improved performancecharacteristics, the technology has several disadvantages. For example,GaAs technology exhibits relatively low yields of properly functioningintegrated circuits. GaAs RF switches tend to be relatively expensive todesign and manufacture. In addition, although GaAs switches exhibitimproved insertion loss characteristics as described above, they mayhave low frequency limitations due to slow states present in the GaAssubstrate. The technology also does not lend itself to high levels ofintegration, which requires that digital control circuitry associatedwith the RF switch be implemented “off chip” from the switch. The lowpower control circuitry associated with the switch has proven difficultto integrate. This is disadvantageous as it both increases the overallsystem cost or manufacture, size and complexity, as well as reducingsystem throughput speeds.

It is therefore desirable to provide an RF switch and method forswitching RF signals having improved performance characteristics.Specifically, it is desirable to provide an RF switch having improvedinsertion loss, isolation, and compression. It is desirable that such anRF switch be easily designed and manufactured, relatively inexpensive tomanufacture, lend itself to high levels of integration, with low-to-highfrequency application. Power control circuitry should be easilyintegrated on-chip together with the switch functions. Such integrationhas been heretofore difficult to achieve using Si and GaAs substrates.The present teachings provide such an RF switch and method for switchingRF signals.

SUMMARY

A novel RF switch circuit and method for switching RF signals isdescribed. The RF switch circuit may be used in wireless applications,and may be fabricated in a silicon-on-insulator technology. In oneembodiment the RF switch is fabricated on an Ultra-Thin-Silicon (“UTSi”)substrate. In one embodiment the RF switch includes: an input forreceiving an RF signal; a first switching transistor grouping connectedto the input to receive the RF signal and connected to an RF commonport, wherein the first switching transistor is controlled by aswitching voltage (SW); a second switching transistor grouping connectedto the first switching transistor grouping and the RF common port,wherein the second switching transistor is controlled by a switchingvoltage SW_, and wherein SW_ is the inverse of SW so that when the firstswitching transistor grouping is on, the second switching transistorgrouping is off. The switching transistor groupings, when enabled,alternatively connect their respective RF input signals to the RF commonport. In this embodiment the RF switch also includes shunting transistorgroupings coupled to the switching transistor groupings and alsocontrolled by the switching voltages SW and SW_. The shunting transistorgroupings, when enabled, act to alternatively shunt their associated RFinput nodes to ground thereby improving RF switch isolation.

The switching and shunting transistor groupings comprise one or moreMOSFET transistors connected together in a “stacked” or serialconfiguration. Within each transistor grouping, the gates of the stackedtransistors are commonly controlled by a switching voltage (SW or SW_)that is coupled to each transistor gate through respective gateresistors. The stacking of transistor grouping devices and gateresistors increases the compression point of the switch. The RC timeconstant formed by the gate resistors and the gate capacitance of theMOSFETs is designed to be much longer than the period of the RF signal,causing the RF voltage to be shared equally across the series connecteddevices. This configuration increases the 1 dB compression point of theRF switch.

A fully integrated RF switch is described that includes digital switchcontrol logic and a negative power supply voltage generator circuitintegrated together with the inventive RF switch. In one embodiment, thefully integrated RF switch provides several functions not present inprior art RF switches. For example, in one embodiment, the fullyintegrated RF switch includes a built-in oscillator that providesclocking input signals to a charge pump circuit, an integrated chargepump circuit that generates the negative power supply voltages requiredby the other RF switch circuits, CMOS logic circuitry that generatescontrol signals to control the RF switch transistors, level-shifting andlow current voltage divider circuits that provide increased reliabilityof the switch devices, and an RF buffer circuit that isolates RF signalenergy from the charge pump and digital control logic circuits. Severalembodiments of the charge pump, level shifting, voltage divider, and RFbuffer circuits are described. The inventive RF switch providesimprovements in insertion loss, switch isolation, and switchcompression. In addition, owing to the higher levels of integration madeavailable by the present inventive RF switch, RF system design andfabrication costs are reduced and reliability is increased using thepresent method and apparatus.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 a is a simplified electrical schematic of a prior art singlepole, single throw (SPST) RF switch used to demonstrate performancecharacteristics of the RF switch.

FIG. 1 b is a simplified electrical schematic of the SPST RF switch ofFIG. 1 a showing the dominant characteristics of the switch when theswitch is turned “on” allowing the RF signal to pass from an input nodeto an output node.

FIG. 1 c shows the equivalent small-signal electrical characteristics ofthe RF switch of FIGS. 1 a and 1 b when the RF switch is turned “off”thereby blocking the RF signal from the output node.

FIG. 2 is a simplified electrical schematic of a prior art single poledouble throw (SPDT) RF switch.

FIG. 3 is an electrical schematic of an RF switch according to oneembodiment of the present method and apparatus.

FIG. 4 is a simplified block diagram of an exemplary fully integrated RFswitch made in accordance with the present method and apparatus.

FIG. 5 a is a simplified block diagram of one exemplary embodiment ofthe negative voltage generator shown in the simplified block diagram ofFIG. 4; FIG. 5 b is an electrical schematic of a first embodiment of acharge pump circuit that is used to generate a negative supply voltageto the RF switch of FIG. 4.

FIG. 5 c is a plot of voltage amplitude versus time showing the voltageamplitude of two non-overlapping clock signals used to control thecharge pump circuit of FIG. 5 b varying over time.

FIG. 6 a is an electrical schematic of a first embodiment of aninventive level shifting circuit; FIG. 6 b is an electrical schematic ofone embodiment of the inverters used to implement the level shiftershown in FIG. 6 a.

FIG. 7 a is a voltage amplitude versus time plot of a digital inputsignal and corresponding output signal generated by the inventive levelshifter of FIG. 6 a; FIG. 7 b is a simplified logic symbol for theinventive level shifter of FIG. 6 a.

FIG. 8 a is an electrical schematic of one embodiment of a two-stagelevel shifter and RF buffer circuit including a first stage levelshifter and a second stage RF buffer circuit; FIG. 8 b is a simplifiedblock diagram of the digital control input and interface to the RFbuffer circuit of FIG. 8 a.

FIG. 9 a is an electrical schematic of one embodiment of a low currentvoltage divider (LCVD) circuit made in accordance with the present RFswitch method and apparatus; FIG. 9 b is a simplified logic symbol usedto represent the voltage divider of FIG. 9 a.

FIG. 10 is an electrical schematic of a second embodiment of a levelshifting circuit using the low current voltage divider circuit of FIG. 9a in combination with the level shifting circuit of FIG. 6 a.

FIGS. 11 a and 11 b are electrical schematics showing an alternativeembodiment of the two-stage level shifter and RF buffer circuit of FIG.8 a.

FIG. 12 is an electrical schematic of a modified charge pump using thelevel shifting circuit of FIG. 10.

Like reference numbers and designations in the various drawings indicatelike elements.

DETAILED DESCRIPTION

Throughout this description, the preferred embodiment and examples shownshould be considered as exemplars, rather than as limitations on thepresent method and apparatus.

The Inventive RF Switch

The present method and apparatus is a novel RF switch design and methodfor switching RF circuits. A first exemplary embodiment of the presentinventive RF switch 30 is shown in FIG. 3. As shown in FIG. 3, in oneembodiment, the inventive RF switch 30 includes four clusters or“groupings” of MOSFET transistors, identified in FIG. 3 as transistorgroupings 33, 34, 37 and 38. Two transistor groupings comprise “pass” or“switching” transistor groupings 33 and 34, and two transistor groupingscomprise shunting transistor groupings 37 and 38. Each transistorgrouping includes one or more MOSFET transistors arranged in a serialconfiguration. For example, in the embodiment shown in FIG. 3, theswitching grouping 33 includes three switching transistors, M_(33A),M_(33B), and M_(33C). Similarly, the switching grouping 34 includesthree switching transistors, M_(34A), M_(34B), and M_(33C). The shuntinggrouping 37 includes three transistors M_(37A), M_(37B), and M_(37C).Similarly, the shunting grouping 38 includes three transistors, M_(38A),M_(38B), and M_(38C). Although the transistor groupings 33, 34, 37 and38 are shown in FIG. 3 as comprising three MOSFET transistors, thoseskilled in the RF switch design arts shall recognize that alternativegrouping configurations can be used without departing from the scope orspirit of the present method and apparatus. For example, as describedbelow in more detail, any convenient number of transistors can be usedto implement the groupings shown in FIG. 3 without departing from thescope of the present method and apparatus.

In one embodiment of the present inventive RF switch, the MOSFETtransistors (e.g., the transistors M_(37A), M_(37B), and M_(37C)) areimplemented using a fully insulating substrate silicon-on-insulator(SOI) technology. More specifically, and as described in more detailhereinbelow, the MOSFET transistors of the inventive RF switch areimplemented using “Ultra-Thin-Silicon (UTSi)” (also referred to hereinas “ultrathin silicon-on-sapphire”) technology. In accordance with UTSimanufacturing methods, the transistors used to implement the inventiveRF switch are formed in an extremely thin layer of silicon in aninsulating sapphire wafer. The fully insulating sapphire substrateenhances the performance characteristics of the inventive RF switch byreducing the deleterious substrate coupling effects associated withnon-insulating and partially insulating substrates. For example,improvements in insertion loss are realized by lowering the transistoron resistances and by reducing parasitic substrate resistances. Inaddition, switch isolation is improved using the fully insulatingsubstrates provided by UTSi technology. Owing to the fully insulatingnature of silicon-on-sapphire technology, the parasitic capacitancebetween the nodes of the RF switch 30 is greatly reduced as comparedwith bulk CMOS and other traditional integrated circuit manufacturingtechnologies. Consequently, the inventive RF switch exhibits improvedswitch isolation as compared with the prior art RF switch designs.

As shown in FIG. 3, similar to the switch described above with referenceto FIG. 2, the transistor groupings are controlled by two controlsignals, SW, and its inverse, SW_. The control signals are coupled tothe gates of their respective transistors through associated andrespective gate resistors. For example, the control signal SW controlsthe operation of the three transistors in the switching transistorgrouping 33 (M_(33A), M_(33B), and M_(33C)) through three associated andrespective gate resistors (R_(33A), R_(33B), and R_(33C), respectively).The control signal SW is input to an input node 33′ to control theswitching transistor grouping 33. SW is also input to an input node 38′to control the shunting transistor grouping 38. Similarly, the inverseof SW, SW_, controls the switching transistor grouping 34 via an inputnode 34′. SW_ is also input to an input node 37′ to control the shuntingtransistor grouping 37.

In one embodiment, the transistor grouping resistors compriseapproximately 30 K ohm resistors, although alternative resistance valuescan be used without departing from the spirit or scope of the presentmethod and apparatus. In addition, in some embodiments of the presentmethod and apparatus, the gate resistors comprise any resistive elementhaving a relatively high resistance value. For example, reversed-biaseddiodes may be used to implement the gate resistors in one embodiment. Asdescribed in more detail below, the gate resistors help to increase theeffective breakdown voltage across the series connected transistors.

The control signals function to control the enabling and disabling ofthe transistor groupings 33, 34, 37 and 38, and the RF switch 30generally functions to pass and block RF signals in a manner that issimilar to the control of the analogous transistors of the switch ofFIG. 2. More specifically, the switching transistor groupings 33 and 34act as pass or switching transistors, and are configured toalternatively couple their associated and respective RF input nodes to acommon RF node 35. For example, when enabled, the switching transistorgrouping 33 couples a first RF input signal “RF₁”, input to a first RFinput node 31, to the RF common node 35. Similarly, when enabled, theswitching transistor grouping 34 couples a second RF input signal “RF₂”,input to a second RF input node 32, to the RF common node 35. Theshunting transistor groupings, 37 and 38, when enabled, act toalternatively shunt their associated and respective RF input nodes toground when their associated RF input nodes are uncoupled from the RFcommon node 35 (i.e., when the switching transistor grouping (33 or 34)that is connected to the associated input node is turned off).

The control voltages are connected to alternatively enable and disableselective pairs of transistor groupings. For example, as shown in FIG.3, when SW is on (in some embodiments this is determined when thecontrol voltage SW is set to a logical “high” voltage level), theswitching transistor grouping 33 is enabled (i.e. all of the transistorsin the grouping 33 are turned on), and its associated shuntingtransistor grouping 38 is also enabled (i.e., all of the transistors inthe grouping 38 are turned on). However, similar to the operation of theswitch of FIG. 2, because the inverse of SW, SW, controls the operationof the second switching transistor grouping 34, and its associatedshunting transistor grouping 37, these two transistors groupings aredisabled (i.e., all of the transistors in the groupings 34. 37 areturned off) during this time period. Therefore, with SW on, the RF₁input signal is coupled to the RF common port 35. The RF₂ input signalis blocked from the RF common port 35 because the switching transistorgrouping 34 is off. The RF₂ input signal is further isolated from the RFcommon port 35 because it is shunted to ground through the enabledshunting transistor grouping 38. As those skilled in the RF switchdesign arts shall recognize, the RF₂ signal is coupled to the RF commonport 35 (and the RF₁ signal is blocked and shunted to ground) in asimilar manner when the SW control signal is off (and the SW_ controlsignal is on).

One purpose of the stacking of MOSFET transistors and using gateresistors as shown in the inventive RF switch 30 of FIG. 3 is toincrease the breakdown voltage across the series connected transistors.The RC time constant formed by the gate resistor and the gatecapacitance of the MOSFETs is designed to be much longer than the periodof the RF signal. Thus, very little RF energy is dissipated through thegate resistor. This arrangement effectively causes the RF voltage to beshared equally across the series connected transistors. The net effectis that the breakdown voltage across the series connected devices isincreased to n times the breakdown voltage of an individual FET, where nis the number of transistors connected in series. This configurationincreases the 1 dB compression point of the inventive RF switch 30.

To achieve improved switch performance, the RC time constant must besized so that it is large with respect to the period of the RF signal.This largely places a constraint on the minimum value of R that can beused to implement the gate transistors. As noted above, in oneembodiment of the present method and apparatus, a typical value of R is30 k-ohms, although other resistance values can be used withoutdeparting from the scope of the present method and apparatus. Because aMOSFET gate input draws no DC current, there is no change in the biasingof the devices due to IR drops across this resistance.

Advantageously, the present inventive RF switch 30 can accommodate inputsignals of increased power levels. Owing to the serial arrangement ofthe MOSFET transistors that comprise the transistor groupings (33, 34,37 and 38), increased power signals can be presented at the RF inputnodes (i.e., at the input nodes 31 and 32) without detrimentallyaffecting switch operation. Those skilled in the transistor design artsart shall recognize that greater input power levels can be accommodatedby increasing the number of transistors per transistor grouping, or byvarying the physical configuration of the transistors. For example, inone embodiment, the transistors are approximately 0.5×2,100 micro-metersin dimension. However, alternative configurations can be used withoutdeparting from the scope or spirit of the present method and apparatus.

Silicon-on-Insulator (SOI) Technologies

As noted above in the description of the RF switch of FIG. 3, SOItechnology is attractive in implementing RF switches due to the fullyinsulating nature of the insulator substrate. As is well known, SOI hasbeen used in the implementation of high performance microelectronicdevices, primarily in applications requiring radiation hardness and highspeed operation. SOI technologies include, for example, SIMOX, bondedwafers having a thin silicon layer bonded to an insulating layer, andsilicon-on-sapphire. In order to achieve the desired switch performancecharacteristics described above, in one embodiment, the inventive RFswitch is fabricated on a sapphire substrate.

Fabrication of devices on an insulating substrate requires that aneffective method for forming silicon CMOS devices on the insulatingsubstrate be used. The advantages of using a composite substratecomprising a monocrystalline semiconductor layer, such as silicon,epitaxially deposited on a supporting insulating substrate, such assapphire, are well-recognized, and can be realized by employing as thesubstrate an insulating material, such as sapphire (Al₂O₃), spinel, orother known highly insulating materials, and providing that theconduction path of any inter-device leakage current must pass throughthe substrate.

An “ideal” silicon-on-insulator wafer can be defined to include acompletely monocrystalline, defect-free silicon layer of sufficientthickness to accommodate the fabrication of active devices therein. Thesilicon layer would be adjacent to an insulating substrate and wouldhave a minimum of crystal lattice discontinuities at thesilicon-insulator interface. Early attempts to fabricate this “ideal”silicon-on-insulator wafer were frustrated by a number of significantproblems, which can be summarized as (1) substantial incursion ofcontaminants into the epitaxially deposited silicon layer, especiallythe p-dopant aluminum, as a consequence of the high temperatures used inthe initial epitaxial silicon deposition and the subsequent annealing ofthe silicon layer to reduce defects therein; and (2) poor crystallinequality of the epitaxial silicon layers when the problematic hightemperatures were avoided or worked around through various implanting,annealing, and/or re-growth schemes.

It has been found that the high quality silicon films suitable fordemanding device applications can be fabricated on sapphire substratesby a method that involves epitaxial deposition of a silicon layer on asapphire substrate, low temperature ion implant to form a buriedamorphous region in the silicon layer, and annealing the composite attemperatures below about 950° C.

Examples of and methods for making such silicon-on-sapphire devices aredescribed in U.S. Pat. No. 5,416,043 (“Minimum charge FET fabricated onan ultrathin silicon on sapphire wafer”); U.S. Pat. No. 5,492,857(“High-frequency wireless communication system on a single ultrathinsilicon on sapphire chip”); U.S. Pat. No. 5,572,040 (“High-frequencywireless communication system on a single ultrathin silicon on sapphirechip”); U.S. Pat. No. 5,596,205 (“High-frequency wireless communicationsystem on a single ultrathin silicon on sapphire chip”); U.S. Pat. No.5,600,169 (“Minimum charge FET fabricated on an ultrathin silicon onsapphire wafer”); U.S. Pat. No. 5,663,570 (“High-frequency wirelesscommunication system on a single ultrathin silicon on sapphire chip”);U.S. Pat. No. 5,861,336 (“High-frequency wireless communication systemon a single ultrathin silicon on sapphire chip”); U.S. Pat. No.5,863,823 (“Self-aligned edge control in silicon on insulator”); U.S.Pat. No. 5,883,396 (“High-frequency wireless communication system on asingle ultrathin silicon on sapphire chip”); U.S. Pat. No. 5,895,957(“Minimum charge FET fabricated on an ultrathin silicon on sapphirewafer”); U.S. Pat. No. 5,920,233 (“Phase locked loop including asampling circuit for reducing spurious side bands”); U.S. Pat. No.5,930,638 (“Method of making a low parasitic resistor on ultrathinsilicon on insulator”); U.S. Pat. No. 5,973,363 (“CMOS circuitry withshortened P-channel length on ultrathin silicon on insulator”); U.S.Pat. No. 5,973,382 (“Capacitor on ultrathin semiconductor oninsulator”); and U.S. Pat. No. 6,057,555 (“High-frequency wirelesscommunication system on a single ultrathin silicon on sapphire chip”).All of these referenced patents are incorporated herein in theirentirety for their teachings on ultrathin silicon-on-sapphire integratedcircuit design and fabrication.

Using the methods described in the patents referenced above, electronicdevices can be formed in an extremely thin layer of silicon on aninsulating synthetic sapphire wafer. The thickness of the silicon layeris typically less than 150 nm. Such an “ultrathin” silicon layermaximizes the advantages of the insulating sapphire substrate and allowsthe integration of multiple functions on a single integrated circuit.Traditional transistor isolation wells required for thick silicon areunnecessary, simplifying transistor processing and increasing circuitdensity. To distinguish these above-referenced methods and devices fromearlier thick-silicon embodiments, they are herein referred tocollectively as “ultrathin silicon-on-sapphire.”

In some preferred embodiments of the method and apparatus, the MOStransistors are formed in ultrathin silicon-on-sapphire wafers by themethods disclosed in U.S. Pat. Nos. 5,416,043; 5,492,857; 5,572,040;5,596,205; 5,600,169; 5,663,570; 5,861,336; 5,863,823; 5,883,396;5,895,957; 5,920,233; 5,930,638; 5,973,363; 5,973,382; and 6,057,555.However, other known methods of fabricating ultrathinsilicon-on-sapphire integrated circuits can be used without departingfrom the spirit or scope of the present method and apparatus.

As described and claimed in these patents, high quality silicon filmssuitable for demanding device applications can be fabricated oninsulating substrates by a method that involves epitaxial deposition ofa silicon layer on an insulating substrate, low temperature ionimplantation to form a buried amorphous region in the silicon layer, andannealing the composite at temperatures below about 950° C. Anyprocessing of the silicon layer which subjects it to temperatures inexcess of approximately 950° C. is performed in an oxidizing ambientenvironment. The thin silicon films in which the transistors are formedtypically have an areal density of electrically active states in regionsnot intentionally doped which is less than approximately 5×10¹¹ cm⁻².

As noted above, UTSi substrates are especially desirable for RFapplications because the fully insulating substrate reduces thedetrimental effects of substrate coupling associated with traditionalsubstrates (i.e., substrates that are not fully insulating).Consequently, in one embodiment, the RF switch 30 of FIG. 3 isfabricated on an UTSi substrate.

RF Switch Design Tradeoffs

Several design parameters and tradeoffs should be considered indesigning and implementing the inventive RF switch 30 described abovewith reference to FIG. 3. The inventive RF switch can be tailored tomeet or exceed desired system design requirements and RF switchperformance objectives. The design tradeoffs and considerations thatimpact the inventive RF switch design are now described.

As described above with reference to FIG. 3, the RF switch 30 isimplemented using MOSFET transistors, which may be “N-type” or “P-type”.However, N channel transistors are preferred for RF switches implementedin CMOS technology. N channel transistors are preferred because, for agiven transistor size, the “on” resistance of an N channel transistor ismuch lower than for a P channel transistor due to the higher mobility insilicon of electrons versus holes. The control voltages are selected toinsure that the on resistance of the “on” transistor is reduced. Thecontrol voltages are also selected to insure that the “off” transistorremains off when disabled.

As is well known in the transistor design arts, in an N channel MOStransistor, the “on” resistance is inversely proportional to thedifference between the voltage applied at the transistor gate and thevoltage applied at the transistor source. This voltage is commonlyreferred to as the “Vgs” (gate-to-source voltage). It is readilyobserved that as the magnitude of the RF signal (Vs) increases at theinput port (e.g., at the first RF input node 31 of FIG. 3), and hence atthe RF common port 35, the Vgs of the on transistors decrease (e.g., theVgs of the transistor M33 _(A) in the switching transistor grouping 33decreases as the magnitude of the RF 1 signal increases). This arguesfor making the gate control voltage (e.g., SW at the input node 33′) aspositive as possible. Unfortunately, reliability concerns limit theextent to which the gate control voltage can be made positive.

A similar concern exists for the “off” transistors. It is important tonote that for typical RF switch applications, the RF input signals(e.g., the RF 1 input signal) generally swing about a zero referencevoltage. The off transistors (e.g., the transistors in the shuntingtransistor grouping 37) must remain disabled or turned off during boththe positive and negative voltage excursions of the RF input signal.This argues for making the gate control voltage of the off transistors(e.g., the SW_ control voltage signal) as negative as possible. Again,reliability concerns limit the extent to which this gate control voltagecan be made negative.

For a CMOS switch, the design of the off transistor also limits the 1 dBcompression point of the switch. As is well known in the transistordesign arts. MOS transistors have a fundamental breakdown voltagebetween their source and drain. When the potential across the deviceexceeds this breakdown voltage, a high current flows between source anddrain even when a gate potential exists that is attempting to keep thetransistor in an off state. Improvements in switch compression can beachieved by increasing the breakdown voltage of the transistors. Onemethod of fabricating a MOS transistor with a high breakdown voltage isto increase the length of the gate. Unfortunately, an increase in gatelength also disadvantageously increases the channel resistance of thedevice thereby increasing the insertion loss of the device. The channelresistance can be decreased by making the device wider, however, thisalso decreases the switch isolation. Hence, tradeoffs exist in MOSswitch designs.

As described above with reference to the inventive RF switch 30 of FIG.3, the transistors are stacked in a series configuration to improve theswitch 1 dB compression point. The relatively high value gate resistors,in combination with the stacking configuration of the transistors in thetransistor groupings, increase the effective breakdown voltage acrossthe series connected transistors. The switch elements are designed andfabricated such that the RC time constant (determined by the resistancevalues of the gate resistors and the gate capacitance of the MOSFETs) ismuch longer than the period of the RF signal processed by the RF switch30. As noted above, the net effect of the stacking configuration and therelatively high resistance gate resistors is to increase the breakdownvoltage across the series connected transistors by a factor of n timesthe breakdown voltage of an individual transistor (where n equals thenumber of transistors connected in series in a transistor grouping).

An additional design consideration concerns the “body tie” used intraditional bulk CMOS transistors. As is well known in the transistordesign arts, the body tie electrically couples the device either to thewell or to the substrate. The well-substrate junction must remainreversed biased at all times. The source-to-body and drain-to-bodyjunctions must remain reversed biased at all times. In general, for bulkCMOS designs, the well (for N-well technology) is tied to the mostpositive potential that will be applied to the circuit. The substrate(for P-well technology) is tied to the most negative potential that willbe applied to the circuit. Because the RF input signal swingssymmetrically above and below ground, bulk CMOS switch designs exhibitpoor insertion loss, isolation, and 1 dB compression point performance.For these reasons, and those described above, the present RF switch 30is preferably implemented on an insulating substrate.

Implementing the inventive RF switch on an insulating substrate providesseveral advantages such as improved switch isolation and reducedinsertion loss. Further advantages are achieved by implementing theinventive RF switch using UTSi technology. For example, as compared withthe prior art RF switch implementations in GaAs, improvements inintegrated circuit yields, reduced fabrication costs, and increasedlevels of integration are achieved using UTSi. As is well known in theintegrated circuit design arts, GaAs does not lend itself to high levelsof integration. Thus, the digital control circuitry and other circuitryassociated with the operation and function of the RF switch (such as anegative voltage power supply generator, level shifting, low currentvoltage divider and RF buffer circuits) must often be implementedoff-chip (i.e., these functions are not easily integrated with the RFswitch). This leads to increased costs and reduced performance of theprior art RF switch implementations.

In contrast, in accordance with the present RF switch method andapparatus, using UTSi technology, the circuitry necessary for the properoperation and functioning of the RF switch can be integrated together onthe same integrated circuit as the switch itself. For example, and asdescribed below in more detail, by implementing the RF switch in UTSitechnology, the RF switch can be integrated in the same integratedcircuit with a negative voltage generator and the CMOS control logiccircuitry required to control the operation of the RF switch. Thecomplexity of the RF switch is also reduced owing to the reduction incontrol lines required to control the operation of the switch.Advantageously, the RF switch control logic can be implemented using lowvoltage CMOS transistors. In addition, even for high power RF switchimplementations, a single, relatively low power external power supplycan be used to power the present inventive RF switch. This feature isadvantageous as compared to the prior art GaAs implementations thatrequire use of a relatively high power external power supply and powergeneration circuitry necessary to generate both positive and negativepower supplies. For example, in the exemplary embodiments describedbelow with reference to FIGS. 4-12, the present inventive RF switchrequires only a single 3 V external power supply. The prior art switchdesigns typically require at least a 6 volt external power supply, andexternal voltage generation circuitry to generate both positive andnegative power supplies.

Fully Integrated RF Switch

FIG. 4 shows a simplified block diagram of an exemplary fully integratedRF switch 100 made in accordance with the present method and apparatus.As shown in FIG. 4, the fully integrated RF switch 100 includes theinventive RF switch 30 described above in FIG. 3 (shown in a simplifiedschematic representation in FIG. 4), CMOS control logic 110, and anegative voltage generator circuit 120 (implemented in one embodimentusing a “charge pump” circuit). A control signal 130 is input to theCMOS logic block 110. In one embodiment, the control signal 130 rangesfrom 0 volts to +Vdd, however those skilled in the digital logic designarts shall recognize that other logic levels can be used withoutdeparting from the scope or spirit of the present method and apparatus.For the reasons provided above, in one exemplary embodiment, the fullyintegrated RF switch 100 is fabricated on UTSi substrates, althoughother insulating substrate technologies can be used.

As described in more detail below, the fully integrated RF switch 100includes several functions and features not present in the prior art RFswitch of FIG. 2. For example, in addition to the inventive RF switch 30(which makes use of the novel transistor stacking and gate transistorconfiguration described above with reference to FIG. 3), the fullyintegrated RF switch 100 integrates the negative voltage generator andRF switch control functions together on the same integrated circuit asthe inventive RF switch. As described below in more detail, the fullyintegrated RF switch 100 includes a built-in oscillator that providesclocking input signals to a charge pump circuit, an integrated chargepump circuit that generates the negative power supply voltages requiredby the other RF switch circuits, CMOS logic circuitry that generates thecontrol signals that control the RF switch transistors, a level-shiftingcircuit that provides increased reliability by reducing thegate-to-drain, gate-to-source, and drain-to-source voltages of theswitch transistors, and an RF buffer circuit that isolates RF signalenergy from the charge pump and digital control logic circuits. Each ofthese circuits is described below in more detail with reference to theirassociated figures.

Negative Voltage Generator—Charge Pump A First Embodiment

As shown in FIG. 4, one embodiment of the fully integrated RF switch 100includes a negative voltage generator or charge pump 120. The negativevoltage generator 120 generates the negative power supply voltage(specified hereafter as “−Vdd”) required by other circuits of the fullyintegrated RF switch 100. Two sets of inputs are provided to thenegative voltage generator 120: a positive DC power supply voltagesignal (Vdd) 122; and a clocking input (shown in the figure as a singleinput signal, “Clk”) 124. Although the clocking input 124 is shown as asingle input signal in FIG. 4, as described below with reference to FIG.5 b, in some embodiments of the present inventive RF switch, theclocking input 124 may comprise two or more clock input signals.

In addition, in the embodiment shown in FIG. 4, the positive supplyvoltage that is input to the negative voltage generator circuit 120comprises a 3 VDC power supply. However, other power supply levels maybe used without departing from the scope or spirit of the present methodand apparatus. For example, if desired, a 3.5 VDC, 5 VDC or any otherconvenient positive DC power supply can be input to the negative voltagegenerator circuit 120 of FIG. 4. The positive power supply signal istypically generated by an external low voltage power supply.

In one embodiment of the present method and apparatus, the negativevoltage generator 120 of FIG. 4 is implemented using a charge pumpcircuit. FIG. 5 a shows a simplified block diagram of one exemplaryembodiment 200 of the negative voltage generator 120 of FIG. 4. As shownin the simplified block diagram of FIG. 5 a, the negative voltagegenerator includes an oscillator 202, a clock generator circuit 204, andan inventive charge pump circuit 206. The oscillator 202 output is inputto the clock generator circuit 204. The output of the clock generatorcircuit 204 is input to the charge pump circuit 206. The negativevoltage generator 120 provides the negative power supply voltage used bythe other circuits of the fully integrated RF switch 100.

Many prior art RF switches disadvantageously require that the negativepower supply voltages be generated by circuitry that is external to theRF switch circuitry. Other RF switch implementations use a couplingapproach necessary to shift the DC value of the RF input signal to themidpoint of the applied bias voltage. This approach generally requiresthat relatively high bias voltages be applied because of the effectivehalving of the FET gate drive due to this level shifting. If the biasvoltages are not increased, this produces a negative effect on theswitch insertion loss because the gate drive is thereby reduced and theFET channel resistances are increased.

To address these problems, one embodiment of the fully integrated RFswitch 100 uses the inventive charge pump circuit 206 shown in detail inFIG. 5 b. As shown in FIG. 5 b, a first embodiment of the charge pumpcircuit 206 includes two P-channel MOSFET transistors, 208 and 210,connected in series with two N-channel MOSFET transistors 212 and 214.The left leg of the charge pump circuit 206 (comprising the firstP-channel transistor 208 connected in series with the first N-channeltransistor 212) is coupled to the right leg of the charge pump circuit(comprising the second P-channel transistor 210 connected in series withthe second N-channel transistor 214) using a first capacitor Cp 216. Thesource of the second P-channel transistor 214 is coupled to a secondcapacitor, an output capacitor, C 218, as shown. Two non-overlappingclock control signals, “Clk1” and “Clk2”, are used to control theoperation of the transistors 208, 210, 212 and 214. For example, asshown in FIG. 5 b, the inverse of “Clk1”, “Clk1_”, control the gates ofthe P-channel transistors 208, 210. The other non-overlapping clockcontrol signal, “Clk2”, controls the gate of the N-channel transistors212, 214, as shown.

The charge pump 206 generates a negative power supply voltage (−Vdd) byalternately charging and discharging the two capacitors (Cp 216 and theoutput capacitor C 218) using the non-overlapping clock input signalsClk1 and Clk2 to drive the transistor gates. The negative power supplyvoltage, −Vdd, is generated from the charge that is stored on thecapacitor C 218. In one embodiment, a pulse shift circuit (not shown) isused to generate a pulse train that drives the charge pump (i.e., thepulse train is input as the clock input signals Clk1 and Clk2). As thepulse train is applied to the charge pump 206, the capacitor Cp 216 isapplied the positive power supply Vdd and then discharged across theoutput capacitor C 218 in an opposite direction to produce the negativepower supply voltage −Vdd. No transistor in the charge pump muststandoff more than Vdd across any source/drain nodes, hence greatlyincreasing the reliability of the charge pump 206.

In one embodiment of the inventive charge pump circuit 206, the output C218 has a capacitance of approximately 200 pF, and Cp 216 has acapacitance of approximately 50 pF. Those skilled in the charge pumpdesign arts shall recognize that other capacitance values can be usedwithout departing from the scope or spirit of the present method andapparatus.

In one embodiment, as shown in the simplified block diagram of FIG. 5 a,the two non-overlapping clock signals are derived from an oscillatorsignal generated by an internal oscillator 202. As shown in FIG. 5 a,the oscillator 202 inputs an oscillation signal to a clock generatorcircuit 204, which in turn, generates the two non-overlapping clocksignals (in any convenient well known manner) that control the chargepump transistor gates. In one embodiment of the present inventive fullyintegrated RF switch 100, the oscillator 202 comprises a relatively lowfrequency (on the order of a few MHz) oscillator. In this embodiment,the oscillator comprises a simple relaxation oscillator. However, asthose skilled in the integrated circuit arts shall recognize, othertypes of oscillators can be used to practice the present method andapparatus without departing from its spirit or scope.

FIG. 5 c shows the voltage amplitude of the two non-overlapping clocksignals, Clk1 and Clk2, varying over time. As shown in FIG. 5 c, the twonon-overlapping clock signals vary in voltage amplitude from −Vdd to+Vdd. In one embodiment, the clock signals vary from −3 VDC to −3 VDC.This arrangement improves the efficiency of the charge pump 206.

The charge pump transistors, 208, 210, 212 and 214 advantageouslycomprise single-threshold N-channel (212, 214) and P-channel (208, 210)devices. Previous charge pump circuits require use of multi-thresholdlevel devices. These previous implementations are therefore more complexin design and cost than the inventive charge pump circuit 206 of FIG. 5b. In one embodiment of the present charge pump 206, the P-channeltransistors 208, 210 have widths of approximately 20 micro-meters, andlengths of approximately 0.8 micro-meters. The N-channel transistors212, 214 have widths of approximately 8 micro-meters, and lengths ofapproximately 0.8 micro-meters. Those skilled in the integrated circuitdesign arts shall recognize that other transistor dimensions can be usedwithout departing from the scope or spirit of the present method andapparatus. The inventive charge pump circuit 206 is very efficient andperforms well despite temperature and process variations.

Level Shifting Circuitry

Because the charge pump circuitry effectively doubles the power supplyvoltages that are applied to the circuit, careful attention must be paidto any potential reliability issues associated with these highervoltages. In order to implement the charge pump in a manner thatincreases the reliability of the transistors, level shifting circuitryis used to limit the gate-to-source, gate-to-drain, and drain-to-sourcevoltages on the transistors to acceptable levels.

An inventive level shifting circuit 300 made in accordance with thepresent method and apparatus is shown in FIG. 6 a. The level shiftingcircuit 300 is used to convert or shift typical or “normal” digitalinput signals (digital signals typically range from ground (GND) to+Vdd) such that they range from −Vdd to +Vdd. The reliability of thefully integrated RF switch transistors is thereby increased. In oneembodiment of the present method and apparatus, the control signals areshifted to −3 VDC to +3VDC, although those skilled in the RF switchcontrol arts shall recognize that other level shifting voltage rangescan be used without departing from the spirit or scope of the presentmethod and apparatus.

As shown in FIG. 6 a, the inventive level shifting circuit 300,hereinafter referred to as the level shifter 300, comprises a pluralityof inverters coupled in a feedback configuration. More specifically, inthe embodiment shown in FIG. 6 a, the level shifter 300 includes twogroups of inverters used to generate first and second shifted outputsignals, “out” on a first output node 314, and its inverse “out_” on asecond output node 316. The first group of inverters comprises inverters302, 304 and 306. A second group of inverters comprises inverters 308,310 and 312. A typical or “normal” digital input signal (i.e., a digitalinput signal that ranges from GND to +Vdd) is input to the level shifter300 at an input node 318 of a first inverter 320. The first inverter 320generates a first input signal “in” (on an output node 324) which isinput to a second inverter 322. The second inverter 322 generates asecond input signal “in_”, the inverse of the first input signal “in”,on an output node 326. Therefore, the first and second inverters, 320,322, generate the signals that are input to the two groups of invertersdescribed above. For example, the first input signal “in” is coupled tothe input 328 of the inverter 302. Similarly, the second input signal“in_” is coupled to the input 330 of the inverter 308.

The output of the first group of inverters, “out”, is generated by afirst output inverter 306, and is provided on a first output node 314.The output of the second group of inverters, “out_”, is generated by asecond output inverter 312, and is provided on a second output node 316.The two level shifter outputs, “out” and “out_”, are input to othercircuits of the fully integrated RF switch 100 of FIG. 4. For example,in one embodiment, the first output, “out”, is coupled to the gates ofthe devices of the switching transistor grouping 33 and the shuntingtransistor grouping 38 (i.e., the “out” signal on the first output node314 of FIG. 6 a is coupled to the “SW” control input signal of FIG. 3,at the input nodes 33′ and 38′, and thereby controls the operation ofthe switching transistor grouping 33 and the shunting transistorgrouping 38 as described above with reference to FIG. 3). Similarly, inthis embodiment, the second level shifter output, “out_”, is coupled tothe “SW_” control input signal of FIG. 3 (at the input nodes 34′ and37′) and thereby controls the switching transistor grouping 34 and theshunting transistor grouping 37 as described above.

The level shifter 300 of FIG. 6 a shifts the DC level of an input signal(i.e., the input signal provided on the input node 318) while leavingthe frequency response of the input signal unchanged. The level shifter300 takes full advantage of the floating technology offered by thesilicon-on-insulator substrate implementation of the fully integrated RFswitch 100. The inverters of the level shifter 300 operate on adifferential basis, i.e., the level shifter shifts the digital inputsignals based upon the difference between two voltage signals. Morespecifically, as long as the difference between the power supply signalsprovided to the inverters (such as, for example, the output inverters306 and 312) is on the order of Vdd, the level shifter 300 reliablyfunctions to shift the input signals to a range between −Vdd to +Vdd. Inone embodiment, Vdd is equal to 3 VDC. In this embodiment, thetransistors comprising the inverters of the level shifter 300 (e.g., theoutput inverters 306 and 312) never have greater than 3 VDC appliedacross their source/drain nodes. This increases the reliability of thetransistor devices.

Referring again to FIG. 6 a, the level shifter uses a feedback approachto shift the digital input signals to voltage levels ranging from −Vddto +Vdd. Specifically, the output of the second group of inverters (308,310, 312) on the second output node 316 (i.e., the “out_” signal) isprovided as feedback to an input of the first group of inverters at theinput of the inverter 304. Similarly, the output of the first group ofinverters (302, 304, 306) on the first output node 314 (i.e., the “out”output signal) is provided as input to the second group of inverters,specifically, is provided as input to the inverter 310.

When the digital input signal on the input node 318 reaches a logical“high” state (i.e., in some embodiments, when the input signaltransitions from GND to +Vdd), the “in” signal (at the node 324) and the“in_” signal (at the node 326) go to ground (e.g., 0 VDC) and Vdd (e.g.3 VDC), respectively. The “out” signal at the first output node 314 isdriven to +Vdd. At the same time, the “out_” signal at the second outputnode 316 is driven towards −Vdd. The feedback (of “out_” fed back to theinput of the inverter 304 and “out” fed forward to the input of theinverter 310) configuration ensures the rapid change in state of thelevel shifter 300. The level shifter 300 works similarly when the inputsignal transitions from a logic high to a logic low state (i.e.,transitions from +Vdd to GND). When the digital input signal on theinput node 318 reaches a logic “low” state, the “in” signal (at the node324) and the “in_” signal (at the node 326) go to Vdd (e.g., 3 VDC), andground, respectively. The “out” signal at the first output node 314 isdriven to −Vdd. At the same time, the “out_” signal at the second outputnode 316 is driven towards +Vdd. The feedback again ensures the rapidchange in state of the level shifter 300. The grounding contributionensures that the level shifter inverters never see more than a full Vddvoltage drop across the source/drain nodes of the MOSFET transistors ofthe inverters.

FIG. 6 b shows one embodiment of the inverters (e.g., the inverters 302,304, and 306) used to implement the level shifter 300 of FIG. 6 a. Asshown in FIG. 6 b, the inverter 340 includes two MOSFET devices, aP-channel transistor 342 and an N-channel transistor 344. The devicesare connected in series as shown, having their gates coupled togetherand controlled by an input signal provided at an input node 346. Thesource of the P-channel transistor 342 is coupled to a first powersupply voltage signal at node 350, while the source of the N-channeltransistor 344 is coupled to a second power supply voltage signal at anode 352. The device drains are coupled together as shown to produce anoutput of the inverter at an output node 348. In one embodiment of thepresent inventive inverter 340, the P-channel transistor 342 has a widthof 5 micro-meters and a length of 0.8 micro-meters. In this embodiment,the N-channel transistor has a width of 2 micro-meters and a length of0.8 micro-meters. Those skilled in the transistor design arts shallrecognize that other physical dimensions can be used for the transistorsof the inverter 340 without departing from the scope or spirit of thepresent method and apparatus. A logical representation of the inverter340 is also shown as symbol 360 in FIG. 6 b.

Thus, using the present inventive level shifter 300, digital inputsignals that initially range from GND to +Vdd are shifted to range from−Vdd to +Vdd. FIG. 7 a shows a voltage amplitude versus time plot of thedigital input signal and the corresponding output signal that isgenerated by the inventive level shifter 300 of FIG. 6 a. As shown inFIG. 7 a, the digital input signal ranges from ground, or 0 VDC to Vdd.The output of the inventive level shifter 300 ranges from −Vdd to +Vdd.In one embodiment of the present inventive RF switch, the input signalranges from 0 VDC to +3 VDC, and the output of the level shifter 300ranges from −3 VDC to +3 VDC. Other values of power supply voltages canbe used without departing from the scope or spirit of the present methodand apparatus. For example, in one embodiment, the input signal canrange from 0 to +3.5 VDC, or from 0 to 4 VDC. In this embodiment, thelevel shifter shifts the signal to range from −3.5 (or −4) VDC, to +3.5(or +4) VDC.

FIG. 7 b shows a simplified logic symbol for the inventive level shifter300 of FIG. 6 a. This logic symbol is used in subsequent figures. Asshown in FIG. 7 b, the digital input signal is provided on the inputnode 318 (the same input node 318 described above with reference to FIG.6 a). The level shifter 300 provides two shifted outputs, “out” and itsinverse “out_”, and these are provided on output nodes 314 and 316,respectively (the same output nodes 314, 316 described above withreference to FIG. 6 a).

RF Buffer Circuit

FIG. 8 a is an electrical schematic of a two-stage level shifter and RFbuffer circuit 400. FIG. 8 b is a simplified block diagram of thedigital control input and interface to the RF buffer circuit 400. Thetwo-stage level shifter and RF buffer circuit 400 of FIG. 8 a comprisesa first stage level shifter 300 and a second stage RF buffer circuit402. The first stage level shifter 300 is identical to that describedabove with reference to FIGS. 6 a, 6 b, 7 a and 7 b, and is thereforenot described in more detail here. As described above, the level shifterstage 300 shifts the logic levels of the digital control signals torange from −Vdd and +Vdd. The second stage of the circuit 400 comprisesthe RF buffer circuit 402. The RF buffer circuit 402 acts as a driverstage only (i.e., no level shifting is performed by the RF buffercircuit).

The RF buffer electrically isolates the digital control signals (such asthose generated by the CMOS logic block 110 of FIG. 4) from the RFswitch 30 described above with reference to FIG. 3. The RF buffer 402functions to inhibit drooping of the control voltages (SW, SW_, whichare also referred to herein and shown in FIG. 8 a as the control signals“out” and “out_, respectively) that control the enabling and disablingof the transistors in the RF switch 30. As described below in moredetail, the RF buffer 402 also functions to prevent coupling of largepower RF signals to the negative power supply (i.e., −Vdd) that isgenerated by the charge pump circuit 206 described above with referenceto FIGS. 5 a-5 c. More specifically, the RF buffer 402 prevents largepower RF signals extent in the RF switch 30 from RF-coupling to, andthereby draining current from, the negative power supply generated bythe charge pump 206 (FIG. 5 b).

When very large power RF input signals are input to the inventive RFswitch 30, coupling of the RF signals to the digital logic signals canoccur unless an RF buffer circuit is used to isolate the digital logicsignals from the RF switch. The RF coupling can and usually willdetrimentally affect the RF transistor control signals (SW and SW_). Forexample, when RF input signals on the order of approximately 30 dBm areinput to a 1 watt RF switch 30, RF coupling can cause voltage swings ofseveral tenths of a volt on the digital control lines. This is due tothe feedback of RF signals from the RF switch through to the digitalcontrol circuitry. This RF coupling effect can adversely affect theenabling and disabling of the RF transistor groupings and hence theproper operation of the RF switch 30. The buffer circuit 402 of FIG. 8 aprevents the undesirable RF coupling effect.

As shown in FIG. 8 a, the inventive buffer circuit 402 is very similarin configuration to the level shifter 300 described above and shown asthe first stage of the two-stage circuit 400. Similar to the levelshifter 300, the RF buffer 402 comprises two groups of inverters, afirst group of inverters (404, 406 and 408) and a second group ofinverters (410, 412, and 414). The output of the first group ofinverters (404, 406, and 408), generated by the first output inverter408, is labeled “out” in the figure and is provided at a first outputnode 416. The output of the second group of inverters (410, 412, and414), generated by the second output inverter 414, is labeled “out_”,and is provided at a second output node 418. The output signal “out_” isthe inverse of the output signal “out”.

Importantly, although the first stage level shifter 300 uses feedback toperform the level shifting function (as described above with referenceto FIG. 6 a), the RF buffer circuit 402 does not feedback its outputsignals to the input. Consequently, the digital input signals input tothe first stage (i.e., the control input signals that are input to thelevel shifter 300 at the nodes 328 and 330) are isolated from the outputsignals that are used to control the RF switch transistors (i.e., thecontrol output signals “out” and its inverse signal “out_” at the outputnodes 416 and 418, respectively, and coupled to the SW and SW_ controlsignal lines, respectively).

More specifically, and referring again to FIG. 8 a, the level shifter300 inputs the digital control signals “in” and its inverse signal “in_”at the nodes 328, 330 respectively (as described in more detail abovewith reference to FIG. 6 a). The first output of the level shifter 300,“out1”, at the output node 314, is fed back to the input of the inverter310 as shown. Similarly, the second output of the level shifter 300,“out1_”, at the output node 316, is fed back to the input of theinverter 304. As described above, because of this feedback topology, RFcoupling occurs (i.e., the level shifter output signals have RF signalssuperimposed thereon) if the output signals of the level shifter areused to directly control the RF switch transistors (i.e., in the absenceof the buffer circuit 402). Therefore the inventive RF buffer circuit402 is used without feedback of the output signals to isolate the inputsignals (i.e., the digital input signals “in” and “in_) from the RFsignals present in the RF switch. As shown in FIG. 8 a, the first outputsignal “out1” of the level shifter 300 is input to the inverters 404,406 of the RF buffer circuit. Similarly, the second output signal“out1_” of the level shifter 300 is input to the inverters 410, 412 ofthe buffer circuit. The two control outputs of the RF buffer circuit 402(“out” and “out_”) control the enabling and disabling of the transistorsof the RF switch and are not provided as feedback to the level shifter.Hence, improved isolation between the RF switch and the digital logiccircuitry is achieved.

In one embodiment, the inverters used to implement the two-stage levelshifter and RF buffer circuit 400 comprise the inverter 340 describedabove with reference to FIG. 6 b. However, those skilled in the inverterdesign arts shall recognize that alternative inverter designs can beused in implementing the two-stage circuit 400 without departing fromthe scope or spirit of the present method and apparatus. In oneembodiment, the transistors used to implement the first stage levelshifter 300 are physically smaller than those used to implement thesecond stage RF buffer circuit 402. Larger dimension transistors areused in the RF buffer circuit 402 to achieve an efficient amplificationof the control signals. For example, in one embodiment, the transistorsused to implement the RF buffer are three times wider than those used toimplement the level shifter 300, resulting in an amplification ofapproximately three times the current. Those skilled in the transistordesign arts shall recognize that other convenient transistor dimensionscan be used to achieve any desired amplification of the digital controlsignals.

Voltage Divider for Use in an Alternative Level Shifting Circuit

FIG. 9 a is an electrical schematic of one embodiment of a low currentvoltage divider (“LCVD”) circuit 500 that is used in the feedback pathof one embodiment of the level shifter 300 described above withreference to FIG. 6 a. FIG. 9 b shows a simplified logic symbol that isused to represent the voltage divider 500 of FIG. 9 a. The voltagedivider 500 is used in one embodiment to address potential gate oxidereliability issues related to excessive voltage swings across the gateoxides of the feedback inverter transistors. As described above withreference to the level shifter 300, although the source-to-drainvoltages of the various MOSFETs used to implement the level shifter arenever applied voltages greater than Vdd, because the outputs of thelevel shifter (i.e., the output signals “out” and “out_) can swing asmuch as 2*Vdd (i.e., from −Vdd to +Vdd), the gate oxides of the feedbackinverters 304 and 310 can have applied voltages of 2*Vdd. These feedbackvoltage levels can be applied across the gate oxides of the feedbackinverters 304, 310, and can result in gate oxide reliability problems.

The gate oxide reliability issues can be averted by ensuring that themaximum voltage applied across the gate oxide of the feedback inverters304, 310 is lowered to approximately Vdd (as contrasted with gate oxidevoltages of 2*Vdd). Therefore, in one embodiment of the presentinventive fully integrated RF switch, the voltage divider of FIG. 9 alimits the voltages applied to the gates of the level shifter feedbackinverters 304, 310. In this embodiment, instead of directly feeding backthe level shifter outputs to their respective feedback inverters asshown in the level shifter of FIG. 6 a (i.e., the outputs “out” and“out_”, at the output nodes 314, 316, respectively), the level shifteroutput signals are first conditioned by the voltage divider 500 of FIG.9 a before being fed back to the feedback inverters. As described belowin more detail, the voltage divider 500 ensures that the voltagesapplied to the gate oxides of the feedback inverters 304, 310 do notexceed more than approximately Vdd plus a small voltage drop (thevoltage drop being a function of the number of transistors used toimplement the voltage divider 500 and a transistor threshold voltage).In one embodiment Vdd is 3 VDC, and the voltage drop is 0.9 VDC. In thisembodiment, the voltage divider 500 ensures that the gate oxides arenever applied voltages exceeding approximately 3.9 VDC (i.e., thefeedback inverters are applied voltages that range from −3 VDC to 0.9VDC).

Referring now to FIG. 9 a, the voltage divider 500 includes a pluralityof MOSFET devices (502, 504, 506 and 508) coupled together in a serialconfiguration (i.e., stacked on top of each other in a source to drainarrangement as shown). In one embodiment, the gate and drain of theMOSFETs 502, 504, 506 and 508 are coupled together to implement stackeddiodes. The diode-implementing MOSFETs, hereafter referred to as “diodedevices”, are stacked in series as shown. The voltage divider 500 alsoincludes a MOSFET M3 510 and an output MOSFET M2 512. The function ofthese two transistors is described in more detail below.

The diode devices are used to divide the voltage of an input signalprovided to the voltage divider 500 at an input node 514. As shown inFIG. 9 a, the signal that is divided by the voltage divider 500 isprovided as input to the drain (and connected gate) of the first device502. Once the input signal exceeds a positive voltage level of (n*Vthn),where “n” is the number of diode devices used to implement the voltagedivider 500, and Vthn is the threshold voltage of the device (i.e., the“diode-drop” from the drain to the source of the device), the diodedevices (502, 504, 506, and 508) begin to conduct current heavily. Inthe embodiment shown in FIG. 9 a, n=4, and Vthn=0.7 volts, althoughalternative values for “n” and Vthn can be used without departing fromthe scope or spirit of the present method and apparatus. For example, inother embodiments, the input signal provided to the divider can belimited to any desired voltage level by varying the number of diodedevices used to implement the voltage divider 500 (i.e., by varying thevalue of “n”). In the embodiment shown in FIG. 9 a, once the inputvoltage exceeds a voltage level of (4*0.7), or 2.8 volts, the stackeddiode devices begin conducting heavily.

A ballast resistor, R 516, is connected to the source of the outputdiode device 508 as shown. Once the diode devices turn on fully, theballast resistor R 516 drops any additional input voltage that exceedsthe value of n*Vthn. In the embodiment shown in FIG. 9 a, the ballastresistor R 516 drops any additional input voltage exceeding the value of(input voltage−(4*Vthn)). The output of the voltage divider 500 istapped from the connected gate-drain of the output diode device 508. Thevoltage-divided output signal is provided on an output node 520. Due tothe diode voltage drops of the diode devices 502, 504, 506, (i.e.,3*Vthn), and the voltage dropped across the ballast resistor R 516, theoutput at the output node 520 is guaranteed to never exceedapproximately (input voltage−(3*Vthn)). For Vthn=approximately 0.7volts, and a maximum input voltage of approximately 3 volts, the outputnode 520 will never exceed (3 VDC−(3*0.7 VDC)), or 0.9 VDC. Thus, in theembodiment shown in FIG. 9 a, for an input voltage ranging between −3VDC to +3 VDC, the voltage divider 500 limits the output of the outputnode 520 to a range of −3 VDC to 0.9 VDC.

The output MOSFET M2 512 is configured as a capacitor and is used toassist in accelerating the switching time of the voltage divider 500.The MOSFET M3 510 assures that the output node 520 swings to thepotential of the input signal at the input node 514 when the input goesto a negative potential. This is accomplished by the device M3 510turning on when the input signal goes to a negative potential. Thus,when the input signal goes to a −Vdd potential (e.g., −3 VDC), theoutput signal at the output node 520 also goes to −Vdd. The outputdevice 508 is reversed biased during negative voltage swings of theinput signal assuring that no DC current is drained from the negativepower supply during the negative voltage swings of the input signal.When the voltage divider output is approximately −3 VDC, the voltagedivider 500 draws no current. This is important because a current at −3VDC discharges the charge pump circuit described above with reference toFIG. 5 b. When the voltage divider output is approximately 0.9 volts,the current that is drawn is very small if the ballast resistor R 516 isselected to be relatively large. However, because the current in thiscase occurs between a positive voltage (0.9 volts) and ground, noadditional charge pump current is delivered due to the operation of thevoltage divider 500 of FIG. 9 a.

In one embodiment, the ballast resistor R 516 has a value of 100 k-ohms.In one embodiment all of the devices of the voltage divider 500 have thesame length. For example, in one embodiment, all of the devices have alength of 0.8 micro-meters. In one embodiment, all of the diode devices(502, 504, 506, and 508) have identical physical dimensions. In oneembodiment, the diode devices each have a width of 2 micro-meters, thedevice M3 510 has the same width of 2 micro-meters, and the outputMOSFET M2 512 has a width of 14 micro-meters. Those skilled in theintegrated circuit design arts shall recognize that other values andalternative configurations for the devices shown in FIG. 9 a can be usedwithout departing from the scope or spirit of the present method andapparatus. For example, those skilled in the electrical circuit designarts shall recognize that other voltage divider output levels can easilybe accommodated by varying the number “n” of diode elements, varying thevalues of Vthn, or by tapping the output node 520 at a different pointin the stack of diode devices (e.g., by tapping the output from thedrain of diode device 506, or 504, instead of from the drain of device508 as shown).

Modified Level Shifter Using the Voltage Divider

By reducing the voltages that are applied to the gate oxides of the RFswitch transistors, the voltage divider 500 of FIGS. 9 a and 9 badvantageously can be used to increase the reliability of thetransistors in both the level shifter 300 and the charge pump circuitdescribed above. For example, FIG. 10 shows a modified level shifter 600using the voltage divider 500 of FIG. 9 a in combination with the levelshifter 300 of FIG. 6 a. As shown in FIG. 10, the output (at output node314) of the inverter 306 of the level shifter 300 is applied to an inputof a first voltage divider 500′. Similarly, the output (at the outputnode 316) of the inverter 312 of the level shifter 300 is applied to aninput of a second voltage divider 500″. The outputs of the voltagedividers are fed back to the input of the feedback inverters 304, 310 asshown in FIG. 10. Specifically, and referring to FIG. 10, the output ofthe first voltage divider, “out”, on the output node 520′ is fed back tothe input of the feedback inverter 310. Similarly, the output of thesecond voltage divider, “out_”, on the output node 520″ is fed back tothe input of the feedback inverter 304. As described above withreference to FIG. 9 a, the level shifters 500′ and 500″ reduce thefeedback voltages to ranges of −Vdd to approximately +0.9 VDC. Thisreduced voltage swing on the feedback paths does not alter the functionof the level shifter 600.

Note that the RF switch control signals, “SW” and “SW_”, can be tappedfrom the level shifter outputs prior to their input to the voltagedividers 500′ and 500″, and provided as input to the inventive RF switch30 of FIG. 3. For example, as shown in FIG. 10, the output of inverter306 at the output node 314 can be tapped and used to generate the switchcontrol signal “SW”. Similarly, the output of the inverter 312 at theoutput node 316 can be tapped and used to generate the switch controlsignal “SW_”. In one embodiment, as described above with reference tothe two-stage level shifter and RF buffer circuit 400 of FIG. 8 a, thecontrol signals tapped from the nodes 314, 316 are first buffered beforebeing coupled to the RF switch transistors. The switch control signals,SW and SW_, are allowed to have a full-rail voltage swing which does notcreate gate oxide reliability problems in the RF switch. Morespecifically, the switch control signals range from −Vdd to +Vdd (i.e.,the voltage levels of the switch control signals are not limited by thevoltage dividers). The full voltage swings of the switch control signalsdo not raise gate oxide reliability issues with respect to the RF switchMOSFETs because the sources of the RF switch MOSFETs are grounded. Theswitch input signals are therefore relative to ground in the RF switchMOSFETs. Consequently, the MOSFETs are applied either a positive Vddvoltage relative to ground across the gate oxides, or a negative Vddvoltage relative to ground across the gate oxides.

FIG. 10 also shows a simplified symbolic representation 601 of a sectionof the modified level shifter 600. The symbol 601 represents the portionindicated by the dashed region 601′ of FIG. 10. As shown in FIG. 10, thesymbolic modified level shifter 601 includes a first input “in_” 630corresponding to the input node 326 (“in_”). The symbolic level shifter601 also includes a second input “out” 632 corresponding to the input tothe feedback inverter 310. Note that this signal is also derived fromthe output 520′ of the first voltage divider 500′. A positive powersupply voltage is input at a +Vdd input 634. A negative power supplyvoltage is input at a −Vdd input 636. The modified level shifter 601 hasthree output signals, “out_pos” (at output 638), “out_neg” (at output640), and “out” (at output 642). These outputs correspond to the outputnodes 606, 608, and 520″ described above. For ease of understanding, thesymbolic representation of the level shifter 601 is used in the figuresdescribed below.

The potential gate oxide reliability problems associated with the levelshifter 300 described above with reference to FIG. 6 a are averted usingthe voltage dividers 500′ and 500″ in the feedback paths of the modifiedlevel shifter 600. In addition, the voltage dividers 500′ and 500″ canalso function to reduce potential gate oxide reliability problemsassociated with the charge pump circuit. As shown in FIG. 10, theoutputs of the inverters 308 and 310 are tapped from the level shifter300 and provided as input to two output inverters to produce two outputsignals, “out_pos” and “out_neg.” More specifically, the output of theinverter 308 is provided as input to a first output inverter 602.Similarly, the output of the feedback inverter 310 is provided as inputto a second output inverter 604.

By coupling the output inverters 602, 604 in this manner, the modifiedlevel shifter 600 output signals never exceed Vdd (or −Vdd). Morespecifically, the first output inverter 602 generates an output signal,“out_pos”, at a first output node 606, that ranges from GND (i.e., 0VDC) to +Vdd. The second output inverter 604 generates a second outputsignal, “out_neg”, at a second output node 608, which ranges from −Vddto GND. When the input signal “in_” goes to GND, the output signal“out_pos” also goes to GND. The output signal “out_neg” transfers fromGND to −Vdd. When the input signal “in_” goes positive to +Vdd,“out_pos” also goes to Vdd, and “out_neg” transfers from −Vdd to GND.Thus, using the present modified level shifter 600, the “out_pos” outputsignal ranges from GND to +Vdd, while the “out_neg” output signal rangesfrom −Vdd to GND. As described below in more detail, the two outputsignals, “out_pos” and “out_neg”, are used to address potential gateoxide reliability problems in a modified charge pump circuit. Asdescribed now with reference to FIGS. 11 a and 11 b, these outputsignals can also be used to address potential gate oxide reliabilityproblems in the RF buffer circuit.

Modified Level Shifter and RF Buffer Circuit

The two-stage level shifter and RF buffer 400 described above withreference to FIG. 8 a can experience voltage swings at the RF bufferinverter inputs of approximately 2*Vdd. As already described, this levelof voltage swing may present gate oxide reliability problems anddetrimentally affect the function of the RF buffer transistors.

FIGS. 11 a and 11 b show an alternative embodiment 400′ of the two-stagelevel shifter and RF buffer circuit 400 described above with referenceto FIG. 8 a. The alternative embodiment of the RF buffer shown in FIG.11 b uses the voltage divider circuit described above to assure thatvoltages on the gate oxides of the RF buffer never exceed greater than0.9 volts above Vdd. As shown in FIG. 11 b, the alternative two-stagelevel shifter and RF buffer circuit 400′ includes a first stage levelshifter circuit 600 coupled to a second stage RF buffer circuit 402′. Inthis embodiment of the level shifter and RF buffer circuit 400′, themodified level shifter outputs, “out_pos” and “out_neg”, described abovewith reference to FIG. 10, are used as input to the RF buffer invertersto generate the RF buffer output signals “out” and “out_”. For example,as shown in FIG. 11 b, the “out_pos” and “out_neg” output signalsgenerated by a first modified level shifter 700 are input to two RFbuffer inverters, 702, 704, respectively. Similarly, the “out_pos” and“out_neg” output signals generated by a second modified level shifter706 are input to two RF buffer inverters, 708, 710, respectively. Inaccordance with the alternative embodiment 400′ shown in FIGS. 11 a and11 b, when an input signal “in” is a logical high signal, the “out_pos”output goes to Vdd while the “out_neg” goes to GND. Thus, when the inputsignal “in” is a logical high value, the output of the inverter 702 goesto GND, and the output of the inverter 704 goes to −Vdd. Therefore, whenthe input signal “in” is high, the output of the inverter 712 (“out”)goes to −Vdd. When the input signal “in” is low, the opposite outputsare produced.

The RF buffer inverters 702, 704 are used to control the power supplyvoltages of a first RF output inverter 712. Similarly, the RF bufferinverters 708, 710 are used to control the power supply voltages of asecond RF output inverter 714. In this embodiment, the RF buffer outputsignals, “out” and “out_”, are used to control the RF switch (i.e.,output signal “out” acts as control voltage “SW”, while “out_” acts ascontrol voltage “SW_”).

Modified Charge Pump An Alternative Embodiment

As noted above, the two output signals “out_pos” and “out_neg” generatedby the modified level shifter 600 of FIG. 10 can be used in analternative embodiment of the charge pump circuit to reduce or eliminatepotential gate oxide reliability problems associated with excessivevoltages applied to the charge pump. As described above with referenceto FIGS. 5 b and 5 c, the clock signals used to control the gates of thecharge pump transistors (i.e., the P-channel transistors 208, 210, andthe N-channel transistors 212, 214) have voltage swings of 2*Vdd. Forexample, as shown in FIG. 5 c, the charge pump clock signals, “Clk1” and“Clk2”, range from the negative power supply voltage−Vdd to the positivepower supply voltage +Vdd. Similar to the gate oxide reliability issuesdescribed above with reference to the RF buffer and level shiftercircuits, this full-rail voltage swing may present oxide reliabilityproblems in the charge pump circuit. Therefore, a modified charge pumpcircuit is shown in FIG. 12 which reduces or eliminates potential gateoxide reliability problems by limiting the voltages applied to gateoxides to range from −Vdd to 0.9 volts.

FIG. 12 shows a modified charge pump 800 that uses the modified levelshifter 600 described above with reference to FIG. 10. As shown in FIG.12, the modified charge pump 800 comprises a charge pump circuit 206′and an inventive charge pump clock generation circuit 802. The chargepump clock generation circuit 802 generates the clock control signalsused by the charge pump circuit 206′. The charge pump circuit 206′ isvery similar in design to the charge pump 206 described above withreference to FIG. 5 b. For example, the charge pump 206′ includes a pairof P-channel transistors 208, 210, and a pair of N-channel transistors212, 214, in addition to a pass capacitor Cp 216 and an output capacitorC 218. In one embodiment of the charge pump circuit 206′, the outputcapacitor C 218 has a capacitance on the order of a few hundred pF, andthe capacitor Cp 216 has a capacitance of approximately 50 pF. Thoseskilled in the charge pump design arts shall recognize that othercapacitance values can be used without departing from the scope orspirit of the present method and apparatus.

The charge pump 206′ functions very similarly to the charge pump 206described above with reference to FIG. 5 a, and therefore its operationis not described in detail again here. The charge pump 206′ shown inFIG. 12 differs from the charge pump 206 in that the control signalsused to control the charge pump 206′ transistor gates (i.e., the gatesof the transistors 208, 210, 212, and 214) are limited to half-railvoltage swings (i.e., they are limited to range from −Vdd to ground, orfrom ground to Vdd). Potential gate oxide reliability problems invokedwhen the gate control voltages are allowed to swing a full rail (i.e.,from −Vdd to Vdd) are thereby reduced or eliminated.

As shown in FIG. 12, the charge pump clock generation circuit 802includes four modified level shifters 804, 806, 808 and 810, coupledtogether in a feedback configuration. In one embodiment of the modifiedcharge pump, the four modified level shifters are implemented by themodified level shifter 600 described above with reference to FIG. 10.FIG. 12 shows the level shifters using the symbolic representation 601of the level shifter 600 of FIG. 10. In this embodiment, the levelshifters 804, 806, 808, and 810 perform identically to the level shifter600 of FIG. 10. The two non-overlapping clock signals, “Clk1”, and“Clk2” (and their inverse signals, “Clk1_” and “Clk2_”, respectively)are input to the “in_” inputs of the level shifters as shown in FIG. 12.The two input clock signals, “Clk1” and “Clk2”, are identical to thenon-overlapping clock signals described above with reference to FIGS. 5a-5 c. As shown above with reference to FIG. 5 c, the twonon-overlapping clock signals vary in voltage amplitude from −Vdd to+Vdd. In one embodiment, the clock signals vary from −3 VDC to +3 VDC.

The four modified level shifters generate the half-rail clock controlsignals that are used to control the charge pump 206′. Specifically, asshown in FIG. 12, the four level shifters generate the “CLK1POS_”,“CLK1NEG_”, “CLK2POS”, and “CLK2NEG” control signals that are input tothe charge pump transistor gate control nodes 250, 252, 254 and 256,respectively. In the embodiment shown in FIG. 12, the level shifters 806and 808 generate the four transistor gate control signals “CLK1POS_”,“CLK1NEG_”, “CLK2POS”, and “CLK2NEG”. The level shifter 806 generatesthe “CLK1POS_” and “CLK1NEG_” gate control signals, while the levelshifter 808 generates the “CLK2POS”, and “CLK2NEG” gate control signals.More specifically, as shown in FIG. 12, the “out_pos” output of thelevel shifter 806 (“CLK1POS_”) is coupled to control the transistor gateinput 250 of the transistor 208. The “out_neg” output of the levelshifter 806 (“CLK1NEG_”) is coupled to control the transistor gate input252 of the transistor 210. Similarly, the “out_pos” output of the levelshifter 808 (“CLK2POS”) is coupled to control the transistor gate input254 of the transistor 214. Finally, the “out_neg” output of the levelshifter 808 (“CLK2NEG”) is coupled to control the transistor gate input256 of the transistor 214. The clock generation circuit 802 functions toprevent excessive voltages across the gate oxides of the charge pumptransistors.

Those skilled in the transistor design arts shall recognize that othercontrol configurations can be used without departing from the spirit orscope of the present method and apparatus. For example, the other twolevel shifters (804, 810) can be used to generate the control signals inan alternative embodiment of the modified charge pump. Also, asdescribed above with reference to the charge pump circuit 206,alternative transistor configurations (N-channel and P-channel) can beused to implement the modified charge pump 206′ of the present methodand apparatus.

As shown in FIG. 12, the four level shifters 804, 806, 808 and 810 arecoupled together in level shifter pairs (804 with 806, and 808 with 810)in a feedback configuration that is very similar to the feedbacktopology of the level shifter described above with reference to FIG. 6a. For example, the “out” output node of the level shifter 804 isprovided as feedback to the “out” node of its associated pair levelshifter 806. Similarly, the “out_” output node of the level shifter 806is provided as feedback to the “out” node of its associated pair levelshifter 804. Similarly, the “out_” output node of the level shifter 808is provided as feedback to the “out” node of its associated pair levelshifter 810. The “out_” output node of the level shifter 810 is providedas feedback to the “out” node of its associated pair level shifter 808.The feedback configuration is used by the clock generation circuit 802in the generation of the four transistor gate control signals“CLK1POS_”, “CLK1NEG_”, “CLK2POS”, and “CLK2NEG”.

Summary

A novel RF switch is provided wherein the switch is fabricated using anSOI CMOS process. Fabricating the switch on an SOI substrate results inlack of substrate bias and allows the integration of key CMOS circuitbuilding blocks with the RF switch elements. Integration of the CMOSbuilding blocks with RF switch elements provides a fully integrated RFswitch solution that requires use of only a single external power supply(i.e., the negative power supply voltage is generated internally by acharge pump circuit integrated with the RF switch). This results inimprovements in RF switch isolation, insertion loss and compression. Inone embodiment, the RF switch has a 1 dB compression point exceedingapproximately 1 Watt, an insertion loss of less than approximately 0.5dB, and switch isolation as high as approximately 40 dB. The inventiveswitch also provides improvements in switching times.

A number of embodiments of the present method and apparatus have beendescribed. Nevertheless, it will be understood that variousmodifications may be made without departing from the spirit and scope ofthe method and apparatus.

Accordingly, it is to be understood that the present method andapparatus is not to be limited by the specific illustrated embodiments,but only by the scope of the appended claims.

What is claimed is:
 1. A circuit, comprising: a) a first RF portconfigured to output or receive a first RF signal; b) a second RF portconfigured to output or receive a second RF signal; c) a switchtransistor grouping having a first node coupled to the first RF port anda second node coupled to the second RF port, wherein the switchtransistor grouping has a control node configured to be coupled to afirst switch control signal (SW); and d) a shunt transistor groupinghaving a first node coupled to the first RF port and a second nodecoupled to ground, wherein the shunt transistor grouping has a controlnode configured to be coupled to a second switch control signal (SW_).2. The circuit of claim 1, wherein the switch transistor groupingcomprises a single MOSFET, and wherein the shunt transistor groupingcomprises a single MOSFET.
 3. The circuit of claim 1, wherein the switchtransistor grouping comprises a plurality of MOSFETs arranged in astacked configuration, and wherein the shunt transistor groupingcomprises a plurality of MOSFETs arranged in a stacked configuration.